Track 8: IBIS / IBIS-AMI Labs - HW Design Masterclass

1. IBIS Fundamentals

What is IBIS?

IBIS (Input/Output Buffer Information Specification) is an industry-standard behavioral model format for describing the electrical characteristics of I/O buffers on integrated circuits. Developed by the Electronic Industries Alliance (EIA), IBIS provides a non-proprietary way to share buffer behavior without exposing the underlying transistor-level circuit design (SPICE netlists).

Unlike SPICE models that describe the physical device physics, IBIS uses measured or simulated I-V (current-voltage) and V-t (voltage-time) lookup tables to capture the essential input/output behavior. This makes IBIS models orders of magnitude faster to simulate while maintaining reasonable accuracy for signal integrity analysis.

Why IBIS Matters

IC vendors provide IBIS models freely because they do not reveal proprietary circuit topology. This enables system-level SI analysis without NDA barriers. Over 90% of commercial SI tools support IBIS natively.

IBIS Model Structure

An IBIS file (.ibs) contains hierarchical data describing one or more components. The key structural elements are:

Section	Purpose	Key Data
`[Component]`	Top-level component definition	Manufacturer, package, pin count
`[Pin]`	Signal-to-pin mapping	Pin name, signal name, model reference
`[Package Model]`	Package parasitics	R_pkg, L_pkg, C_pkg per pin
`[Model]`	I/O buffer behavior	Model type, I-V curves, V-t waveforms
`[Ramp]`	Slew rate fallback	dV/dt_r, dV/dt_f
`[Rising Waveform]`	Accurate transition shape	Time-voltage pairs under specific load
`[Falling Waveform]`	Accurate transition shape	Time-voltage pairs under specific load

I-V Curves

The heart of an IBIS model is its set of I-V (current-voltage) characteristic tables. These describe the static (DC) behavior of the buffer's output transistors and protection structures. There are four fundamental I-V tables:

Pullup I-V: Current from VDD through the pull-up structure (PMOS in CMOS) as a function of voltage at the pad. Defined relative to the VDD rail.
Pulldown I-V: Current from the pad through the pull-down structure (NMOS in CMOS) to VSS. Defined relative to GND.
Power Clamp I-V: Current through the ESD protection diode connected to VDD. Active during overshoot events.
GND Clamp I-V: Current through the ESD protection diode connected to GND. Active during undershoot events.

IBIS I-V Table Format

[Pulldown]
| Voltage(V) | I_typical(A) | I_min(A) | I_max(A) |
| -1.0 | -50.0m | -45.0m | -55.0m |
| 0.0 | 0.0 | 0.0 | 0.0 |
| 1.8 | 24.0m | 21.0m | 27.0m |
| 3.6 | 26.0m | 23.0m | 29.0m |

V-t Curves (Waveform Tables)

V-t waveform tables capture the dynamic (transient) behavior of the output buffer. They record the voltage at the output pad versus time during a rising or falling transition under a specific test load. Typically, two load conditions are provided:

Test fixture 1: R_fixture = 50 ohm to VDD (for falling waveform) or GND (for rising waveform)
Test fixture 2: R_fixture = 50 ohm to GND (for falling waveform) or VDD (for rising waveform)

The simulator interpolates or extrapolates from these waveforms to predict behavior under different loading conditions. This is why at least two load conditions are needed for each transition direction.

Three Corners: Typ/Min/Max

Every IBIS data table provides three columns representing silicon process, voltage, and temperature (PVT) corners:

TYPICAL

Nominal process, nominal voltage (e.g., 1.8V), 25C. Represents expected production average behavior.

MIN/MAX CORNERS

Min = slow/weak driver (slow process, low voltage, high temp). Max = fast/strong driver (fast process, high voltage, low temp). Must simulate both to capture worst-case behavior.

IBIS File Syntax Deep Dive

An IBIS file uses a plain-text format with keywords enclosed in square brackets. The hierarchical structure begins with file-level metadata, followed by component definitions, and finally model definitions. Understanding the syntax is essential for reading and debugging IBIS files.

IBIS File Header Keywords ▼

Keyword	Required	Description
`[IBIS Ver]`	Yes	IBIS specification version (e.g., 5.0, 6.1, 7.0)
`[File Name]`	Yes	Name of the .ibs file
`[File Rev]`	Yes	Revision number of this file
`[Date]`	No	Creation/modification date
`[Source]`	No	How the data was generated (measured, simulated)
`[Notes]`	No	Free-form notes about the model
`[Disclaimer]`	No	Legal disclaimer from the IC vendor
`[Copyright]`	No	Copyright information

Model Type Keywords ▼

The [Model] section's Model_type keyword determines the buffer topology:

Model_type	Buffer Type	Active Structures
`Output`	Push-pull driver	Pullup + Pulldown + both clamps
`Input`	Receiver input	Clamps only (high-Z)
`I/O`	Bidirectional	Selectable Output or Input mode
`3-state`	Tri-state driver	Output mode or high-Z mode
`Open_drain`	Open drain output	Pulldown + GND clamp only
`Open_source`	Open source output	Pullup + Power clamp only
`I/O_open_drain`	Bidirectional open drain	Pulldown + both clamps
`Series`	Series switch element	On/Off resistance tables

IBIS Model Validation

Before using an IBIS model for signal integrity simulation, it must be validated. The IBIS Open Forum provides the official ibischk tool (now at version 7.x) that performs syntax and semantic checking. Common validation errors include:

Non-monotonic I-V tables: Current values must increase monotonically with voltage in pulldown tables
Insufficient voltage range: I-V tables must cover at least -VDD to 2*VDD
Missing corners: All three columns (typ/min/max) must be populated
Waveform time alignment: Rising and falling waveforms must start near the pre-transition steady state
Negative resistance regions: Can indicate measurement or extraction errors

Best Practice: Always Validate Before Use

Run ibischk on every IBIS file before using it in simulation. A "Pass" result does not guarantee accuracy, but a "Fail" or "Warning" often reveals data quality issues that would produce incorrect simulation results. Many SI tools include built-in IBIS validators.

IBIS Version History and Feature Evolution

Version	Year	Key Addition
IBIS 1.0	1993	Basic I-V, V-t, package RLC
IBIS 2.1	1994	Multi-source specification, power clamp refinements
IBIS 3.2	1999	Series elements, bus hold, submodels
IBIS 4.0	2002	Package matrix models, multi-port package
IBIS 5.0	2008	IBIS-AMI (Algorithmic Modeling Interface) for SerDes
IBIS 5.1	2012	Enhanced AMI, backchannel support
IBIS 6.0	2013	BIRD (Buffer Issue Resolution Document) process
IBIS 6.1	2015	Enhanced multi-die packages, 3D-IC support
IBIS 7.0	2019	Unified IBIS-AMI flow, enhanced measurement specs
IBIS 7.1	2021	PAM4 support, enhanced jitter modeling

2. Driver/Receiver Modeling

IBIS Driver Model Architecture

An IBIS output (driver) model consists of several parallel current paths from the pad to the supply rails. The following SVG diagram illustrates the canonical IBIS driver model topology:

The IBIS driver model combines static I-V tables for pullup/pulldown transistors and ESD clamp diodes with V-t waveform tables for transient behavior. Package parasitics (R, L, C) connect the die pad to the external pin.

Pull-up / Pull-down I-V Tables

The pullup I-V table describes current flow from VDD through the PMOS (or equivalent) transistor to the pad as a function of pad voltage. Convention: voltage is measured relative to VDD. When the pad is at VDD, current is zero. As the pad drops below VDD, the PMOS conducts, sourcing current to the pad.

The pulldown I-V table describes current from the pad through the NMOS (or equivalent) transistor to GND. Voltage is measured relative to GND. When the pad is at 0V, current is zero. As pad voltage increases, the NMOS sinks current from the pad.

During a logic transition, the simulator switches between pullup-active (driving high) and pulldown-active (driving low) states. In the input or high-impedance state, both are off, and only the clamp diodes remain active.

Clamp Diodes

The Power Clamp and GND Clamp I-V tables model the ESD protection diodes present on every I/O pin:

Power Clamp: Conducts when pad voltage exceeds VDD + Vf (diode forward voltage). Protects against overshoot. Current flows from pad to VDD rail.
GND Clamp: Conducts when pad voltage drops below GND - Vf. Protects against undershoot. Current flows from GND rail to pad.

Clamp currents are always present regardless of driver state (active driving or high-Z). They are critical for modeling overshoot/undershoot clamping behavior accurately.

Common IBIS Modeling Error

Missing or incorrect clamp data can lead to simulations that show unrealistically large overshoot/undershoot. Always verify that power and ground clamp tables are populated in the IBIS file. Some vendors provide "zero clamp" entries which effectively disable clamp modeling.

Rising and Falling Waveforms

The [Rising Waveform] and [Falling Waveform] sections contain time-voltage pairs that precisely describe the output transition shape under specific test loads. These are more accurate than the [Ramp] section which only provides a linear dV/dt approximation.

Each waveform is measured with a specific test fixture (R_fixture, V_fixture, C_fixture). The simulator uses at least two waveforms (different loads) to derive the buffer's composite behavior through interpolation. The accuracy of the simulated waveform depends heavily on these tables.

Waveform Table Example

[Rising Waveform]
R_fixture = 50.0
V_fixture = 0.0
| time(s) | V_typical(V) | V_min(V) | V_max(V) |
| 0.000e-9 | 0.050 | 0.060 | 0.040 |
| 0.200e-9 | 0.120 | 0.140 | 0.100 |
| 0.500e-9 | 0.850 | 0.900 | 0.780 |
| 1.000e-9 | 1.650 | 1.700 | 1.580 |
| 1.500e-9 | 1.780 | 1.790 | 1.760 |

3. Package Parasitics

R-L-C Package Model

Every IC pin connects the silicon die pad to the external package pin through a physical path that includes bondwires, lead frame traces, or solder bumps. This path has parasitic resistance (R), inductance (L), and capacitance (C) that affect signal integrity.

Parameter	Typical Range	Dominant Effect	Frequency Impact
R_pkg (Resistance)	0.01 - 0.5 ohm	DC voltage drop, damping	Low frequency
L_pkg (Inductance)	0.5 - 10 nH	di/dt noise, ringing, SSO	High frequency (dominant)
C_pkg (Capacitance)	0.1 - 5 pF	Added load, frequency limit	High frequency

Package Types and Parasitics

BGA (Ball Grid Array)

L_pkg: 0.5 - 3 nH (short solder bump path)
Advantage: Minimal bondwire inductance, short interconnect length, excellent for high-speed I/O
Used in: DDR4/5 memory, high-speed FPGAs, processors

QFP (Quad Flat Pack)

L_pkg: 3 - 15 nH (long bondwire + lead frame)
Disadvantage: High inductance from bondwires (1-3mm length), significant SSO noise
Limited to: Lower-speed interfaces, cost-sensitive designs

Bondwire Inductance

Bondwire inductance is often the dominant package parasitic. A typical gold bondwire has approximately 1 nH per millimeter of length. The inductance creates voltage noise proportional to di/dt during switching:

Bondwire Noise Voltage

V_noise = L_bondwire * (di/dt)

For L = 3 nH, di = 50 mA, dt = 0.5 ns:
V_noise = 3e-9 * (50e-3 / 0.5e-9) = 0.30 V

SSO (Simultaneous Switching Output) Noise

When N output drivers switch simultaneously, the total di/dt through shared package inductance causes ground bounce: V_bounce = N * L_shared * (di_per_pin/dt). For 16 pins switching through a 2 nH shared ground bondwire, the bounce can exceed 1V, causing false triggering on neighboring signals.

Interactive Package Delay Calculator

Package Propagation Delay & Impedance

Package Inductance (nH):

Package Capacitance (pF):

Package Resistance (ohm):

Propagation Delay

54.8 ps

Package Z0

54.8 ohm

Resonant Freq

2.91 GHz

Advanced Package Models: Coupled RLC Matrix

For high-pin-count BGA packages, the simple per-pin R/L/C model is insufficient. The IBIS specification supports a coupled package model defined as an N-port RLC matrix that captures mutual coupling between adjacent pins. This is essential for accurately predicting crosstalk between pins sharing the same package substrate.

Coupled Package Model Matrix (4-pin example)

4. IBIS Waveform Simulations

Understanding how different IBIS model parameters affect the output waveform is critical for predicting signal quality. The following interactive simulation lets you adjust key driver parameters and observe their impact on the output waveform at the receiver.

Driver Output Waveform Simulator

Observations: Higher output impedance reduces overshoot but slows the signal. Faster slew rate causes more ringing. More load capacitance degrades rise time. The Max corner (fast process, high VDD, low temperature) produces the most aggressive waveform.

How Different IBIS Models Produce Different Waveforms

Different IC vendors and different I/O standards produce markedly different IBIS model characteristics. Key differentiators include:

I/O Standard	Typical Z_out	Slew Rate	VDD	Overshoot Tendency
LVCMOS 3.3V	25-40 ohm	2-4 V/ns	3.3V	Moderate (unterminated)
LVCMOS 1.8V	30-50 ohm	1-3 V/ns	1.8V	Low to moderate
LVDS	~100 ohm diff	~1 V/ns	3.3V	Very low (diff pair)
SSTL-15 (DDR3)	~40 ohm	3-5 V/ns	1.5V	Controlled (ODT)
HSTL-12 (DDR4)	~34/40 ohm	4-6 V/ns	1.2V	Controlled (ODT)
POD-12 (DDR5)	~34 ohm	5-8 V/ns	1.1V	Low (pseudo open-drain)

Multi-Model Comparison View

Compare two I/O standard models side-by-side. The trace length adds propagation delay and interconnect effects. Notice how different driver strengths and slew rates produce different settling behaviors at the receiver.

Waveform Quality Metrics

When comparing IBIS model waveforms, several quantitative metrics help characterize signal quality:

Metric	Definition	Good Value	Concern Threshold
Rise Time (10-90%)	Time for signal to transition from 10% to 90% of VDD	< 2x minimum spec	> 3x minimum spec
Overshoot %	(V_peak - VDD) / VDD * 100	< 10%	> 20%
Undershoot %	(V_min / VDD) * 100	> -10%	< -20%
Settling Time	Time to reach within 5% of final value	< 2 ns	> 5 ns
Ring Count	Number of overshoots before settling	0-1 rings	> 3 rings
Monotonicity	Signal reaches threshold only once	Monotonic	Non-monotonic

Effect of Termination Strategy

The termination scheme dramatically affects the received waveform, particularly for unterminated LVCMOS interfaces versus terminated DDR/SSTL interfaces:

UNTERMINATED (LVCMOS)

No termination resistor. Reflection coefficient = +1 at receiver (open circuit). Signal doubles at receiver end, then bounces between driver and receiver. Overshoot can exceed 100% of VDD on the first reflection. Multiple rings before settling. Acceptable only for short traces (< 2 inches for 100 MHz signals).

PARALLEL TERMINATED (SSTL/DDR)

50 ohm termination to VTT at receiver. Reflection coefficient near zero (matched impedance). Clean single transition with minimal overshoot (< 5%). No ringing. Required for DDR and any interface above 200 MHz. Power consumption from DC bias current through termination resistor.

IBIS Model Corner Analysis Best Practice

Production silicon varies across process, voltage, and temperature (PVT). A robust SI analysis must simulate all three IBIS corners (Typ/Min/Max) and verify that timing and voltage margins are met across all corners. The worst cases are often non-obvious:

Setup time worst case: Fast driver (Max corner) + long trace + slow receiver (Min corner)
Hold time worst case: Slow driver (Min corner) + short trace + fast receiver (Max corner)
Overshoot worst case: Fast driver (Max corner) + long unterminated trace
Undershoot worst case: Fast driver (Max corner) + impedance mismatch at receiver

5. IBIS-AMI Equalization Concepts

What is IBIS-AMI?

IBIS-AMI (Algorithmic Modeling Interface) extends the standard IBIS model to support SerDes (Serializer/Deserializer) channel simulation. While basic IBIS models describe static I-V behavior and waveforms, IBIS-AMI adds the ability to model the complex equalization algorithms used in multi-gigabit serial links.

Modern SerDes operate at data rates from 5 Gbps to over 100 Gbps per lane. At these speeds, the channel (PCB traces, connectors, cables) introduces severe inter-symbol interference (ISI) that degrades the signal eye. Equalization is essential to "open" the eye at the receiver. IBIS-AMI models capture these equalization algorithms as compiled dynamic libraries (DLL on Windows, SO on Linux).

Two-Pass AMI Simulation

IBIS-AMI simulation uses a unique two-pass approach: (1) The analog channel impulse response is convolved through the entire TX-channel-RX path. (2) The AMI DLLs process the bit stream to add equalization effects. This separates linear (channel) from non-linear (equalization) processing for efficiency.

TX Equalization: Pre-emphasis and De-emphasis

The transmitter uses FIR (Finite Impulse Response) filtering to pre-compensate for channel loss. There are two equivalent perspectives:

Technique	Mechanism	Effect	Use Case
Pre-emphasis	Boost amplitude on transitions	Increases high-frequency content at transitions, compensates channel roll-off	Short-to-medium reach links
De-emphasis	Reduce amplitude on non-transitions	Reduces low-frequency content at sustained bits, same spectral shaping as pre-emphasis	Preferred interpretation (lower peak power)
Pre-cursor EQ	FIR tap at t-1	Compensates for pre-cursor ISI from reflections	Channels with reflections
Post-cursor EQ	FIR taps at t+1, t+2...	Compensates for post-cursor ISI from channel loss	Most common use case

TX FIR Equalizer (3-tap)

V_out[n] = C(-1) * V_in[n+1] + C(0) * V_in[n] + C(1) * V_in[n-1]

Where:
C(-1) = Pre-cursor tap coefficient (typically -0.1 to 0)
C(0) = Main cursor tap (typically 0.7 to 1.0)
C(1) = Post-cursor tap (typically -0.3 to 0)
Constraint: |C(-1)| + |C(0)| + |C(1)| = 1.0

RX Equalization: CTLE and DFE

The receiver applies two main equalization stages to recover the data signal:

CTLE (Continuous Time Linear Equalizer)

CTLE is an analog filter that boosts high-frequency content to compensate for channel loss. It operates on the continuous analog signal before sampling. Key characteristics:

Linear filter (can be represented by transfer function)
Boosts high frequencies while passing low frequencies
Amplifies noise along with signal (noise penalty)
Typical peaking: 0 to 15 dB at Nyquist frequency
Modeled in AMI as a configurable analog EQ stage

DFE (Decision Feedback Equalizer)

DFE is a non-linear digital equalizer that uses previously decided bits to cancel post-cursor ISI. Unlike CTLE, DFE does not amplify noise:

Non-linear (uses hard bit decisions, not representable as transfer function)
Subtracts ISI from known previous bits
Does not amplify noise (major advantage over CTLE)
Typical configurations: 1-tap to 12-tap DFE
First tap is most critical (largest post-cursor)

Interactive Eye Diagram: Before/After Equalization

Eye Height: -- mV | Eye Width: -- ps | Status: Adjust equalization to open the eye

AMI Parameter Files (.ami)

Every IBIS-AMI model comes with a parameter file (.ami) that defines the user-configurable equalization settings. The .ami file uses a Lisp-like syntax with nested parentheses to define parameter trees:

Example .ami Parameter File Structure

(my_serdes_tx
  (Description "High-speed SerDes TX model")
  (Type "Output")
  (tx_tap_units
    (Usage Info)
    (Type Integer)
    (Description "Number of TX FIR taps")
    (Default 3)
  )
  (tx_precursor
    (Usage In)
    (Type Float)
    (Range -0.2 0.0 -0.1)
    (Default -0.05)
    (Description "Pre-cursor tap coefficient")
  )
  (tx_postcursor
    (Usage In)
    (Type Float)
    (Range -0.3 0.0 -0.15)
    (Default -0.1)
  )
)

Channel Operating Margin (COM)

COM is an IEEE 802.3-defined figure of merit that predicts whether a channel meets a target BER (typically 10^-12 or 10^-15). COM combines the effects of channel loss, equalization, noise, and jitter into a single dB value. A COM > 3 dB indicates the link has sufficient margin.

COM (Channel Operating Margin)

COM = 20 * log10( A_signal / A_noise )

Where:
A_signal = Equalized eye height at the sampling point
A_noise = Total noise (ISI residual + crosstalk + random noise) at target BER

COM > 3 dB: Compliant link (sufficient margin)
COM > 0 dB: Marginal (may pass under ideal conditions)
COM < 0 dB: Non-compliant (will fail)

Equalization Adaptation Algorithms

Modern SerDes automatically adapt equalization settings during link training. The key adaptation algorithms include:

Algorithm	What It Adapts	Convergence Time	Complexity
LMS (Least Mean Squares)	DFE tap coefficients	~1000 UI	Low
SS-LMS (Sign-Sign LMS)	DFE taps (simplified)	~5000 UI	Very Low
Zero-Forcing	CTLE + DFE settings	~500 UI	Medium
MMSE (Min Mean Square Error)	All EQ parameters	~2000 UI	High
Backplane EQ Protocol	TX presets via backchannel	~10K UI	Protocol-dependent

AMI Model Quality Checklist

When evaluating an AMI model from a vendor, verify: (1) Correlation data vs silicon measurements is provided, (2) Model supports both statistical (GetWave=False) and time-domain (GetWave=True) analysis, (3) Jitter and noise parameters are properly defined, (4) The model converges within a reasonable number of UI during adaptation.

6. Interactive Labs

Lab 6.1: Model Parameter Tuning

Adjust driver strength and slew rate to meet timing requirements while keeping overshoot within specification.

Rise Time: -- ns | Overshoot: --% | Settling Time: -- ns | Adjust parameters to meet specs

Lab Goal

Target: Rise time < 1.5 ns, Overshoot < 15%, Settling time < 3 ns. Find the optimal combination of driver strength and slew control for a given trace length.

Lab 6.2: Channel Simulation (Driver to Receiver)

Simulate a complete signal path from driver through PCB channel to receiver. Observe how channel characteristics affect the received signal.

Observe waveforms at driver output (blue), mid-channel (yellow), and receiver input (green). Termination strategy dramatically affects reflection behavior.

Lab 6.3: Signal Optimization - Tune EQ to Open Eye

You have a 25 Gbps NRZ link with a lossy channel. Tune the TX and RX equalization to achieve a minimum eye opening of 40 mV height and 20 ps width.

Eye Height: -- mV | Eye Width: -- ps | Target: > 40 mV height, > 20 ps width | Tune EQ to open the eye

Lab Tips: Common IBIS Simulation Pitfalls

Pitfall 1: Incorrect Package Model

Many IBIS files contain a "default" package model with generic R/L/C values that may not match the actual package used. Always verify the [Package Model] section matches your specific component package (QFP, BGA, CSP, etc.). Using wrong package parasitics can shift the resonant frequency by 50% or more, completely invalidating your simulation results.

Pitfall 2: Corner Mismatch

Ensure the simulation tool is using the correct corner (Typ/Min/Max). Some tools default to "Typical" only. For compliance analysis, you must simulate all three corners and verify worst-case behavior. The min-max range can show 30-50% variation in rise time and 2x variation in overshoot.

Pitfall 3: ODT Modeling

On-Die Termination (ODT) for DDR interfaces requires careful setup in the IBIS model. Many DDR IBIS files provide multiple [Model Selector] entries for different ODT settings (RZQ/6, RZQ/7, etc.). Using the wrong ODT value will produce incorrect waveform and timing results. Always match the ODT setting to your memory controller configuration.

Industry Tools for IBIS/AMI Simulation

Tool	Vendor	Strengths	Typical Use Case
HyperLynx	Siemens EDA	Integrated with PCB design flow, easy IBIS setup	Pre-layout/post-layout SI analysis
SIwave + Circuit	Ansys	Full 3D EM extraction + IBIS-AMI simulation	Complex multi-board systems
ADS	Keysight	Industry reference for AMI, extensive SerDes library	SerDes compliance verification
Questa SI	Siemens EDA	Statistical eye, batch AMI analysis	DDR and SerDes optimization
SPISim	SPISim	Fast IBIS/AMI validation, COM analysis	Model QA, quick compliance check
PyBERT	Open Source	Free, Python-based AMI analysis	Learning, initial exploration

7. Knowledge Check - IBIS / IBIS-AMI

Q1: What is the primary advantage of IBIS models over SPICE transistor-level models for I/O buffer simulation?

They provide more accurate results at all frequencies They are non-proprietary and simulate much faster while protecting IP They can model power supply noise more accurately They include thermal simulation capabilities

IBIS models are behavioral (table-based) models that do not reveal transistor-level IP. They simulate orders of magnitude faster than SPICE while providing sufficient accuracy for signal integrity analysis.

Q2: In an IBIS model, what do the Power Clamp and GND Clamp I-V tables represent?

The main driver transistors when actively driving The pull-up and pull-down resistance values ESD protection diodes that clamp overshoot and undershoot The package parasitic model elements

Power and GND clamp tables model the ESD protection diodes that conduct during overshoot (voltage above VDD) and undershoot (voltage below GND) events. They are always active regardless of driver state.

Q3: Why does an IBIS model require at least two waveform tables (different load conditions) for each transition direction?

To represent minimum and maximum process corners To allow the simulator to interpolate/extrapolate the waveform under arbitrary loading conditions To separately model rising and falling edge behavior To validate the I-V table data consistency

Two load conditions are needed so the simulator can determine how the buffer's output waveform changes with different loads. With only one waveform, the simulator cannot predict behavior under a different load. Interpolation between two known load points enables reasonable accuracy across a range of loads.

Q4: What is the dominant package parasitic that causes SSO (Simultaneous Switching Output) noise?

Package capacitance (C_pkg) Package inductance (L_pkg), especially shared ground/power bondwires Package resistance (R_pkg) Inter-pin mutual capacitance

SSO noise is caused by V = L * di/dt, where L is the package inductance (primarily bondwire inductance) shared by multiple output drivers. When N pins switch simultaneously, the total di/dt through the shared inductance creates a noise voltage that appears as ground bounce or power sag.

Q5: What is the key advantage of DFE (Decision Feedback Equalizer) over CTLE (Continuous Time Linear Equalizer)?

DFE provides more equalization gain at high frequencies DFE operates on the analog signal before the CDR DFE cancels ISI without amplifying noise DFE consumes less power than CTLE

DFE uses previously decided (hard) bit values to subtract known ISI from the current sample. Since it subtracts a known quantity rather than boosting the signal, it does not amplify noise. CTLE, being a linear filter, boosts both signal and noise at high frequencies.

Q6: In a 3-tap TX FIR equalizer with coefficients C(-1)=-0.1, C(0)=0.7, C(1)=-0.2, what type of equalization is being applied?

Pure pre-emphasis with no de-emphasis Pure DFE equalization Combined pre-cursor and post-cursor de-emphasis CTLE peaking at Nyquist frequency

C(-1) = -0.1 is the pre-cursor tap (compensates pre-cursor ISI), C(0) = 0.7 is the main cursor, and C(1) = -0.2 is the post-cursor tap (compensates post-cursor ISI). The negative pre and post taps create de-emphasis on sustained bits, boosting transitions. The sum of absolute values = 1.0 as required.